Understand the basics of current and voltage feedback op amps and where they fit, along with constraints and implications
With both current and voltage feedback operational amplifiers (op amp) available to the system designer, how do you decide which one to use? This article is the second of two parts. It reviews several applications that are most suitable to the current feedback type, then it introduces the fully differential amplifier (FDA). All applications covered in Part 1 for voltage feedback op amps are also suitable for an FDA, but some particularly useful applications for this type of amplifier are shown here.
Keep in mind that applications requiring flexibility in the gain setting network will benefit from the gain bandwidth independence of the current feedback (CFB) design. As described in Part 1, the loop gain equation for a CFB includes only the feedback impedance while the gain element can be varied freely with minimal bandwidth interaction. The following examples exploit that advantage in several situations where a CFB would be the preferred solution.
If an inverting summing amplifier is required with a signal bandwidth that is independent of the number of channels summed or the required gains in those channels, a CFB amplifier should be used. Figure 1 shows an example of this approach using the high output voltage and current THS3091.
Here, we assume a 0 Ω source for each of the signal sources where each channel would see a gain of "2 V/V. For these CFB designs, the feedback resistor is first picked to be close to the recommended value for that particular CFB amplifier. Then, each input resistor should be selected according to the gain required by that channel.
Recall that this circuit implemented with a voltage feedback (VFB) has a bandwidth set by the noise gain (NG). For instance, five channels summed with a gain of -2 V/V will have a noise gain of 11. This condition would set a bandwidth for all five channels reduced to the [gain bandwidth product (GBP)]/11 even though each signal only sees a gain of -2 V/V. Using a CFB for this application retains the bandwidth much better because the parallel combination of all gain resistors does not enter strongly into the loop gain equation.
By way of comparison, Figure 2 simulates a single channel of Figure 1 using first the CFB TH3091, then a similar VFB THS4031. The THS4031 is a low-noise, high-voltage VFB that offers approximately 200 MHz GBP.
The circuit of Figure 1 produces a noise gain of 11, which shows up as a signal bandwidth close to 18 MHz for the THS4031, while the THS3091 gives about 200 MHz for each channel.
Where frequency response peaking is required, a CFB amplifier permits this characteristic to be achieved with reduced interaction between the gain shaping and the amplifier bandwidth. Figure 3 shows an example single stage of a zero/pole pair using a high output current OPA691 CFB amplifier
This example transitions from a gain of 2 V/V (6 dB) to a gain of +20 V/V (26 dB) over a 2 MHz to 20 MHz span. Implementing this design with a VFB requires a minimum gain bandwidth product (GBP) in excess of (20 × 20 MHz) = 400 MHz in order to not immediately roll off at the maximum gain setting.
Figure 4 shows this simulation where the improved performance of the CFB OPA691 is apparent.
As a comparison, Figure 4 also shows the high output current OPA690 VFB; note that the 300 MHz GBP of the OPA690 is not quite enough for this application while the CFB OPA691 achieves the maximum 26 dB gain at 20 MHz and remains there up to an approximate -3 dB bandwidth of 100 MHz.
Sallen-Key, or voltage-controlled voltage source, active filters need a non-inverting gain amplifier that has a bandwidth far in excess of the desired filter bandwidth. While this type of filter can be implemented with VFB devices quite well, a CFB device would be preferred where higher frequency cutoff filters are needed, or where the amplifier gain needs to be flexible. The amplifier gain enters into the ideal filter transfer function as part of the Q setting equation. This gain also sets the low frequency gain in a low-pass filter design.
The local bandwidth of the op amp used in this design moves the actual filter poles away from the design targets. In the Sallen-Key low-pass filter, the actual poles move down and to the right in the complex s-plane. This shift gives an actual filter that has lower 0 and higher Q than targeted. To the extent that the amplifier bandwidth changes with gain setting, as it would with a VFB amplifier, the actual filter poles are impacted more strongly using a VFB over a range of gains in the design.
Using a CFB in this filter normally allows a more solid pole placement where the amplifier gain can be varied more freely in the design process because there will be minimal interaction between the required amplifier gain and the impact on the final filter pole locations.
Figure 5 shows an example 20 MHz low-pass Butterworth filter implemented using the CFB OPA695 that gives a passband gain of +4 V/V.
The resistor values have been adjusted slightly from an ideal analysis to account for the amplifier bandwidth effects and hit the desired frequency response exactly.
In this design, the OPA695 gives an amplifier bandwidth >400 MHz at a gain of +4 V/V (12 dB). This bandwidth gives a 20x margin to the desired filter poles. It is this bandwidth margin that allows the filter poles to be implemented with minimal production variation and holds the stop band rejection down to higher frequencies. All Sallen-Key filters show an increasing gain (or reduced stop band rejection) at very high frequencies as the closed loop output impedance of the amplifier increases sufficiently to support a feed through path through the filter feedback capacitor.
As a comparison, Figure 6 shows the filter of Figure 5 simulated first with the OPA695 and then with the VFB OPA820. Operating at a gain of +4 V/V, the OPA820 will have a bandwidth of about 80 MHz.
The OPA695 implementation hits the desired maximally flat Butterworth design with a 20 MHz cutoff almost exactly. The OPA820 placed into the same circuit shows a slight peaking and reduced bandwidth as predicted. This performance contrasts with the MFB filter discussed in Part 1 of this article, where a VFB is preferred and the effect of finite amplifier gain bandwidth product (GBP) moves the actual poles on a constant 0 circle to lower Q.
Where the gain needs to be adjusted, a CFB amplifier is preferred because it holds more constant bandwidth as that adjustment is made. Figure 7 shows an example non-inverting design where the adjustment is configured to provide a fine-scale tune from a gain of +2 to +4 V/V using a high output current OPA691.
In all of these CFB circuits, the feedback resistor is selected and fixed near the recommended value for that particular CFB device. Any adjustments or frequency response shaping is then done using only the gain element.
Since the loop gain does not depend strongly on the signal gain, this type of adjustment holds a more constant bandwidth using a CFB as opposed to a VFB implementation. Conversely, this circuit with the gain adjustment in the feedback resistor would have a significant frequency response variation when using a CFB.
To compare, Figure 8 shows the circuit of Figure 7 simulated at the gain extremes using both the CFB OPA691 and a very similar VFB device, the OPA690.
At a gain of +2 V/V (6 dB), both parts show about the same bandwidth (280 MHz) where the OPA691 is closer to a Butterworth response while the OPA690 is bit more peaked. Adjusted to a gain of +4 V/V (12 dB), the OPA691 still holds >220 MHz bandwidth while the OPA690 drops to around 100 MHz.
Most simple circuits not mentioned thus far can generally use either a VFB or CFB device. It is sometimes suggested that some of these circuits cannot be implemented using a CFB device; this claim is often incorrect. One example that demonstrates this is the differencing amplifier.
A single amplifier differential stage (sometimes called a 'differencing' amplifier) can use either VFB or CFB devices. The common-mode rejection ratio (CMRR) of a CFB implementation appears to be lower when compared to an equivalent VFB design. However, that CMRR is the effect of the buffer gain loss across the input stage and is quite repeatable for a particular CFB part number. It is possible to tune up the CMRR to a much higher dc level for a CFB differencing amplifier by slightly increasing the resistor to ground on the non-inverting input.
Figure 9 shows a representative differencing amplifier using the very high slew rate, unity gain stable, VFB OPA690. Notice also that the resistors on the non-inverting side were adjusted down to achieve a matched input impedance for each source (if V1 and V2 are independent sources. See Reference 1 for an example of where they are not).
A series resistor into the non-inverting input is then added to achieve bias current cancellation, which would only work to improve output offset voltage in a VFB implementation. This is assuming 0 Ω sources for each source and two independent sources.
This same circuit can be built using the CFB OPA691. Figure 10 shows a CMRR simulation where both inputs are tied together and driven.
The resulting small output gain (large negative dB gain) is then input-referred and the negative taken to get a typical CMRR plot. The OPA690 shows slightly higher CMRR than the OPA691. In this case, the resistors of Figure 9 have been used in both simulations, and no adjustment for improved CMRR made in the OPA691 simulation.
Differential input/output circuits
An emerging class of amplifiers called fully differential amplifiers (FDA) easily can take a single or differential input signal. and produce a differential output centered on a user-selected common-mode operation point. An alternative approach in going differential-in to differential-out has been to use standard dual op amps. A brief review of that approach helps to set the background for the FDA. These approaches are useful also because once they are understood, they open up a large range of dual op amps to the designer for possible application.
Differential I/O circuits can be easily implemented using either a VFB or CFB. There is, however, some difference between a non-inverting or inverting input implementation with regard to how the common-mode voltage is treated. In the non-inverting input case, the two inputs show a high input impedance to the differential source (allowing filters or other passive circuits to be easily inserted up to these inputs). The common-mode gain from the non-inverting inputs to the differential outputs will be one.
Figure 11 shows an example of this design where the wideband, high output current, dual OPA2614 is used to implement a DSL driver.
Here, the amplifier operates on a single +12 V supply and the filtered differential source is driven through blocking capacitors to the mid-supply referenced termination impedance. The differential gain is set to 6 V/V with a dc blocking capacitor in series with the gain resistor to further attenuate low-frequency noise and reduce output differential offset voltage.
Whether that capacitor is present in the circuit or not, the mid-supply reference on each non-inverting input shows up as the output common mode voltage as well. This circuit demonstrates a typical 1:2 step-up transformer at the output to a 100 Ω load, giving a nominal differential load of 50 ohms.
This circuit is often implemented using dual current feedback op amps (such as the OPA2677) but can give lower output noise using a decompensated dual voltage feedback op amp (for example, the OPA2614). Figure 12 shows the simulated frequency response for this circuit showing about 50 MHz bandwidth—more than adequate for most DSL line driver applications.
In the inverting input, differential I/O implementation, the differential input impedance is the sum of the two gain resistors; the output common-mode voltage depends on the dc voltage applied to the non-inverting inputs and the dc gain for that signal path, along with the dc common-mode voltage of the source. If the sources are capacitively or transformer-coupled, the common-mode voltage applied to the non-inverting inputs will have a gain of one to the output.
Figure 13 shows an inverting differential I/O using the very low power OPA2684 dual CFB in a single +5 V supply with a mid-supply common-mode reference and an ac-coupled input interface.
This circuit provides a common-mode output of 2.5 V with a differential gain of five and >100 MHz bandwidth while using only 3.6 mA total quiescent supply current.
Figure 13 also shows a differential ac input impedance of 400 ohms. One advantage of the OPA2684 is that the source impedance (possibly a filter) will not interact with the amplifier bandwidth. A dual VFB amplifier used here will work, but the source impedance would then be part of the loop gain equation and might adversely impact the frequency response.
Figure 14 illustrates a simulation of this low-power inverting input differential I/O example where a dual VFB device, the OPA2822, is also shown.
That part is a unity gain stable, 200 MHz GBP device so this noise gain of six configuration shows about 35 MHz bandwidth versus >150 MHz bandwidth for the OPA2684.
Fully differential amplifiers are essentially VFB devices with an added output common-mode control loop. Instead of an internal differential to single-ended conversion, such as standard op amps require, these devices continue the signal path to the output differentially. All of the applications discussed in Part 1 for VFB devices would also be suitable for an FDA device adapted to a differential signal path. However, there are at least two types of application circuits where the FDA provides a compelling solution as compared to standard VFB implementations.
A dc-coupled, single-ended input to differential output with output common-mode control can best be implemented with an FDA device. One of the key considerations in this design is to match the feedback divider ratios on each side of the FDA circuit, including the signal source impedance. It is also important to understand that the common-mode control loop is level-shifted from the input to the output by setting up a common-mode current in the feedback paths. Therefore, the source must be able to sink or source a portion of this dc common-mode current. (See Reference 2 for a discussion of basic FDA operation and applications.)
Figure 15 shows an example circuit using the THS4511, a very wideband, single, 5V-supply FDA that includes ground in its input common-mode range.
This feature makes the THS4511 particularly useful for converting a single-ended, ground-referenced signal that swings only above ground into a differential output around a common-mode voltage.
The THS4511 shows a very high full-power bandwidth with its 2 GHz GBP and 4900 V/μsec differential slew rate. These two characteristics together give an exceptional pulse response in a dc-coupled single to differential conversion. Figure 16 shows the simulated frequency response.
Differential-input-to-differential-output circuits with very low distortion can clearly benefit from the FDA topology. Where a low IF requirement needs the best third-order intermodulation spurious suppression using modest quiescent power levels, a transformer-coupled FDA implementation provides a surprisingly low-noise figure with exceptionally low harmonics.
Figure 17 shows an example of this type of circuit using the very wideband THS4509.
This example gives a noise figure of 8.2 dB (from a 50 Ω source) while also giving >100 dB two-tone spurious free dynamic range (SFDR) through 70 MHz for 2 Vpp outputs. This performance is equivalent to a 54 dBm third-order intercept in an FDA using approximately 200 mW quiescent power.
The 1:1.4 input turns-ratio transformer reflects the 100 Ω differential input impedance of the FDA circuit to a 50 Ω termination (Reference 3 discusses this circuit and measured performance in detail). In this case, the transformer becomes the bandwidth limiting element. Figure 18 shows the simulated performance, including the transformer model.
This single to differential gain of 7 V/V shows up as a low frequency gain of 16.9 dB. The first rolloff above is the transformer while the second break in the rolloff curve is the appearance of the THS4509 bandwidth limitation. As can be seen, this circuit gives very good flatness through 200 MHz IF frequencies.
Today's circuit designers enjoy a tremendous selection of high performance wideband op amps. Newer and increasingly better devices are emerging steadily, showing constant improvement on the speed versus power tradeoff. Where the feedback element needs to be adjustable or is capacitive, a voltage feedback or fully differential device is the preferred solution. Where gain flexibility or frequency response shaping is desired in a low-power implementation, and dc precision is a secondary concern, a current feedback device would be the first choice. Many applications can use either a VFB or CFB device where issues such as the speed/power tradeoff, noise or dc precision become the deciding factor.